Tag Archives: precision attenuation measurement

HPAK 8763B switch test

The 8763B is a very useful device, a 4-port coaxial switch, and has been sold for many years by HP, Agilent, and still sold by Keysight today.
It is single-ended terminated, and has two latching switches.

8763b switch

Two of these will give a nice transfer switch, for auto-calibration (through-connection vs. DUT) of the attenuator test setup.

These switches are specified up to 18 GHz, and have a “max. 0.03 dB insertion loss repeatability”. Now, the big question is, what is the actual repeatability. Knowing the manufacturer, it can be 10 or 100 times better, but you never know. This is fairly critical, because a combined uncertainty of 0.06 dB, for the two 8763B forming the full transfer switch would be not acceptable for the purpose of calibrating attenuators to better than 0.1 dB precision/linearity.

So, quickly hooked this up to the not yet auto-calibrating setup, and recorded power traces, 40 points each, 1 measurement per second, and switching the 8763B in and out every 10 seconds (vertical lines).

This was done at 4, 8, 12, and 18 GHz, and for all ports of the switch.

The setup
8763b test setup
(green item on the right hand side is the feed line directional coupler, connected to the Micro-Tel SG-811 source; light blue test cable on the left is going to the Micro-Tel 1295 receiver).

The results (two examples; same finding at all frequencies) are not very difficult to interpret:

8763b test 1

8763b test 2

– There is not really any switching visible, and one can only judge that the repeatability is actually +-0.002 to +-0.004 – the noise of the measurement.

It seems the only way to get more accuarate data will be to measure the repeatability with the two switches in series, in the final setup. Even though I’m using high quality microwave test cables, 0.002 dB amplitude stability, at 18 GHz is a challenge.
Will need to let the source and receiver fully warm up and stabilize, and use long integration times, like several minutes per switching event, to get data of 0.001-0.002 dB resolution. For now, it seems the switches will add much less uncertainty to the setup as initally thought.

Attenuator calibration – first real dataset!

Some items of the mighty precision attenuator calibrator setup are still missing, like the automatic auto-zero/through calibration, and the adaption of the reflection bridge (see earlier post), but nevertheless, all parts are now in place to do some first real measurements (and generate, thanks to computer control, more data than anyone could have ever recorded manually, without getting close to nervous breakdown).

The device unter test (DUT). It is a HPAK 33321 SG step attenuator, 35 dB, in 5 dB steps – it is more or less a transmission line, with 3 resistor pads that can be individually switched in and out.
33321 sg step attenuator DUT

Also, note the SMA to SMA connector needed to get things connected vs. a through line. No allowance was made for this connector, it is a 18 GHz qualified precision part and will have very little loss.
hp 33321 sg data
As you can see, it is specified for 4 GHz operation – there are multiple models of these attenuators, both from Weinschel/Aeroflex and HP/Agilent/Keysight, up to 26.5 GHz or more. The 4 GHz models are relatively easily to come by, and others have claimed that these are actually quite usable up to higher frequencies. Let’s see.

While I don’t have exact specs for the 33321 SG model, there are similar models around, and typically, the numbers given by HP are +-0.2 dB at 10 dB, +-0.4 dB at 20, and +-0.5-0.7 in the 30-40 dB range. Repeatability about +-0.01 dB, which is quite impressive.

To be exact, we will be dealing with insertion loss here – not quite attenuation, but close, because no corrections have been made for any return losses (due to the SWR of the receiver and of the DUT, which might also change with attenuation setting).

Now, the test:

Step (1) – the system was calibrated without the DUT, just with the cables (from generator and to receiver) directly coupled (“through” calibration)
Step (2) – the attenuator was inserted, and tested at all the steps from 0 to 35 dB, 0 dB was measured twice. For all steps, 10 readings were taken, 1 per second, and averaged. Standard deviations are very small, showing the excellent short-term stability of the setup:
sdev vs frq at various attenuations

Step (3) – again, a through calibration. The measurements took about 3 hours – the drift was small, and distributed linearly with time over the measurements. Drift is pretty much independent of frequency. Later, there will be a drift correction anyway, by the yet-to-be-implemented auto-calibration feature.

Drift – 3 hours; about 0.1 dB absolute per hour.
drift in dB, 3 hours period

Insertion loss – at all the various steps, relative to “through” connection
insertion loss vs. through connection

Insertion loss – relative to “0 dB” setting. This is relevant for most of the practical cases, where the 0 dB values are stored in a calibration ROM, and the levels corrected accordingly. Repeatability of the 0 dB setting was also checked – standard deviation is about 0.04 dB, but might be much better short-term (more like the 0.01 dB claimed in the datasheet). However, keep in mind, 0.04 dB at 10-18 GHz is not more than a few mm of cable movement, or a little bit more or less torque on a connector.

insertion loss (corrected for 0 dB loss) vs frequency

Deviation of 0 dB corrected loss from the nominal values (5-10-15-20-25-30-35 dB steps)
total insertion loss - 35 db to 0 db, 5 db steps - deviation actual-nominal

As you can see, the attenuator works great well above the 4 GHz, easily to 12 GHz. Still usable up to 18, with some error. This error seems to mainly come from the 20 dB pad. Rather than relying on just the 20 dB pad measurement, some maths were done on the data to determine the insertion loss difference (attenuator switched in, vs. switched out), for each off the pads, e.g., for the 20 dB pad, by subtractions of these measurements:

(1) 20 dB in, 5 and 10 out; vs 0 dB
(2) 5 and 20 in, vs 5
(3) 10 and 20 in, vs 10

So there are actually 3 combinations for each pad that allow determiation of the actual insertion loss, for each individual pad. Furthermore, this utilizes the 1295 received at different ranges of the (bolometer) log amplifier, and with different IF attenuators inserted – and will average out any slight errors of the log amp, and calibration errors of the IF step attenuators of the 1295. For even more cancelation, the source power could be varied, but fair enough.

Results of a lot of (computerized) number crunching:

Insertion loss difference in vs. out, for each pad
attenuation of each pad vs frequency
The 5 and 10 dB pads are performing great, the 20 dB pad – a bit less. Well, there must be a way to tune this a bit – but don’t have a cleanroom here, and the fixtures, to scratch a bit of resistive mass from the pad, at certain places, etc. Wonder how they do this at the factory, and if in fact there is some manual tuning, at least for the higher frequency units.

Deviation from nominal, for each pad
calculated attenuation for each of the pads - actual minus nominal

This is really a quite striking level of accuracy – much better than specification, and also indicates the level of precision already achievable with the still temporary attenuation calibration setup. Up to 12 GHz, no issues at all.
33321 sg pad in vs out values

The 0 dB loss – some might be in the connectors, some in the transmission lines, some in the “through” switches of the 3 attenuators. Simply assuming that there aren’t any losses in the connectors and transmission lines, this is the loss per attenuator switch, when in “through”=”pad switched out” position.
0 dB insertion loss (calculated), per contact

All in all, the best way to use these attenuators obviously is to very accurately measure the 0 dB insertion loss, on a pretty narrowly spaced frequency scale. For the attenuator pads, these are best measured by recording values at various attenuations, and polynomial fits give very good approximation, without the need for a lot of density on the frequency scale, and seem to be merely additive, with little cross-correlation errors.
Sure, such things can all be analyzed with much more maths involved, but I doubt it will impact much the application-relevant aspects, and would be rather just a numerical exercise.

Micro-Tel 1295 receiver: parallel to serial converter – digital readout

The Micro-Tel 1295 has a GPIB (IEEE-488) interface, and in principle, can be fully controlled through this. In principle, but, not with ease; and, as it turns out, the build-in processor is running on 70s hardware, and doesn’t respond well to my National Instrument GPIB card. The only thing I need are the attenuation readings, in dB, same as shown on the front LED display.

Also, these GPIB cards are expensive, and I would rather like to control the whole attenuator calibration rig through one single USB port – also to be able to run in with various computers, not just with a dedicated machine.

In brief, after trying hard, I gave up – there need to be a more practical way to read the 1295 data.

First, how to get the data out, if not through the IEEE-488 interface? The case if fully closed, and drilling a hole, mounting a connector – NO. The modification should be reversible.
But there is a solution – the band selection connector, which is already used to remotely control the band switching, has a few spare pins!

This connector is a sight by itself:
micro-tel 1295 band control (and now serial output) plug amphenol wpi mini hex series 126

AMPHENOL/Wire-Pro “WPI” 9-pin “126 series” miniature hexagonal connector, 126-220; these connectors have been introduced in the 1940s, or latest, in the 50s. Still, available today… but the first piece of test equipment that I have ever seen that uses such kind of connectors. 500 Volts, 7.5 Amp – seems like a lot for such a small connector, at 14.99 USD each (plug only).

So, how to run the full display info over one or two wires? Update rate is 1 reading per second, or 1 reading every 4 seconds, not a lot of data – still it needs to be reliable, easy to use.
After some consideration, I decided to use a RS232 interface, with TTL level logic (rather than RS232 voltage levels-only using a short cable), and running it at 2400 bps, transmitting the data from the 1295 receiver to the main micro. This main controller, an ATmega32L can easily handle one more incoming signal, via its USART, and buffer any data coming from the 1295 before it is requested over the USB bus, by the PC software.

There are 5 full digits, plus a leading 1, a optional “+”, a leading-0 removing signal, and a blanking signal, which is set to low when the display is updating, or when the receiver is not giving a valid reading (over/underrange). Each digit needs 4 bits, binary coded decimal (BCD), so in total: 5×4+4=24 bits. Perfect match for the A, B, and C ports of a ATMega32L. This micro will monitor the blanking signal, and after sensing an updated display, read out the BCD information, convert it into a readable string, and send it out at the 2400 bps, via a single wire, no handshake, or anything. Will just keep on sending.

The easiest way to get the signal was determined to be directly at the display unit (A7) itself.
1295 a6 assembly schematic

There is also some space to fit the micro board, a commercial (Chinese, called “JY-MCU”, Version 1.4) ATMega32L minium board, with USB connection. These are really great, running at 16 MHz, with some little LEDs (which are on port B – disabled for this application), and a bootloader. It just saves a lot of time, and these boards are really cheap, below $10.

A 34 pin ribbon cable, with double row connector, salvaged from an old PC – so the controller/parallel-to-serial converter can be removed from the 1295 if no longer needed, and even the cable de-soldered.

The modified assembly
1295 a6 assembly top
MC14511 CMOS latches-BCD decoders-LED drivers – very common for 70s/early 80s vintage, and still working great!

1295 a6 assembly

Data is being transmitted, no doubt:
2400 bps signal

Now it is just a matter of some lines of code, and soon, some real insertion loss tests can start! Stay tuned.

PLL characterization – final results for the Micro-Tel SG-811 and Micro-Tel 1295 circuits

After some experimentation, measurements, etc. – as described before, time to wrap it up.

The PLL loop filter output is now connected to the phase lock input (the additional 1 k/100 n low pass in the earlier schematic has been omitted), with a 330 Ohm resistor in series. This will remain in the circuit, because it’s handy to characterize the loop, and to provide a bit of protection for the opamp output, in case something goes wrong, to give it a chance to survive.

With the charge pump current adjustments now implemented in the software, that’s the result, all pretty stable and constant over the full range.

The SG-811 signal source
micro-tel sg-811 pll bandwith vs frequency

The 1295 receiver
micro-tel 1295 pll bandwidth vs frequency

Micro-Tel SG-811 PLL: frequency response
Gain
sg-811 final gain

Phase
sg-811 final phase

Micro-Tel 1295: frequency response
Gain
1295 pll final gain

Phase
1295 pll final phase

Noise and spurs, ADF41020/Micro-Tel SG-811 PLL

After getting things worked out with the loop filter, some quick check for spurious responses. To do such analysis near the noise level, a FFT/dynamic signal analyzer can be used, but I find it somewhat troublesome, and rather use a swept frequency analyzer for any such work that goes beyond 1 kHz. Below 1 kHz, the FFT is hard to beat. One of the few exemptions is the HPAK 3585A spectrum analyzer, which covers from about DC to 40 MHz, and has resolution bandwidth filters of down to 3 Hz (discrete hardware, not software filters), with baseline at -135 dBm, or lower.

The 3585A doing its thing…
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The results – 1 to 500 Hz
348_00_0001 to 05
Mainly 60 Hz harmonics – well, will need to keep the cables short (especially the coarse tune cables) and everything far away from mains transformers.

10 Hz to 5 kHz
348_00_001 to 5
Signal at 1 kHz is about -70 dBm, not much. No spurs.

5 to 30 kHz
348_00_5 to 30
Two unexpected spurs – 1st: 19.986 – this is an artifact of the 3585A. 2nd: 18766, this seems unreleated to the PLL (doesn’t change with frequency or divider settings), maybe some switchmode supply stray. Well, down below -100 dBm.

25 kHz (with some 60 Hz harmonic sidebands, -115 dBm) – reference spur, about -93 dBm.
25 khz spur detail

All in all, with some refinement of the software, and a bit of mechanical work to get this all mounted into a nice case, the setup should work find and provide great service.
Sure enough, some direct phase noise measurements on the SG-811 output will eventually follow, once the opportunity is right and the equipment at hand.

Micro-Tel SG-811/ADF41020 PLL: working out the details – loop filter, bandwidth, charge pump currents

Designing a stable PLL is not really a big challenge, with all the simulation tools available, and after you have mastered some basic experiments with the 4046 chip, or similar circuits. For PLL simulation software, I suggest to look at ADIsimPLL, available free of charge, from Analog Devices.
However, stable doesn’t necessarily mean wideband, and exhibiting similar characteristics over a full 2 to 18 GHz band. That’s what we want to achieve here.

First some targets – after reviewing the circuits of the Micro-Tel SG-811/1295, and looking at the stability of the build-in YIGs, I figured that a good PLL bandwidth for this system would be somewhere in the 200-500 Hz region. This would still allow to correct for some mains-induced frequency fluctuations (50/60 Hz), and the frequencies are well below the 25 kHz phase detector frequency used for the ADF41020. Furthermore, the bandwidth should be reasonably stable of the full range of frequencies, with no need to use multiple loop filters, or troublesome switchable capacitors/variable gain amplifiers – all should be operated from a single-ended 15V power supply, to provide 0-10 V for the Micro-Tel 1295, and 0-3 V for the SG-811, from a single little board.

With this in mind, an OPA284 rail-to-rail precision amplifier (low noise, 4 MHz BW, can drive +-6.5 mA) was selected as the active part, and some capacitors (only use good quality capacitors, polymer dielectric, or stable ceramic capacitors, NPO) and resistors put together. There is only one adjustment, the damping resistor in the feedback loop.

Sketch of the schematic
adf41020 sg-811 pll loop filter

How to figure out the loop characteristics? Many pages have been written about this, determining open-loop gains and phase margins, etc., but how to approach this in practice, one you have done the calculations and figured out a setup that basically works? This is where the extra resistor and the two test points (A, B, see schematic) come into play. The resistor close to the output (8k2, this is just a temporary part, only inserted during test – bridged with a piece of view during normal operation) is used to isolate the loop output, from the SG-811 phase lock input (which is nothing else than a heavy VCO=voltage controlled oscillator). A few extra parts are also connected to feed a test signal to the VCO, in addition to the loop filter output voltage.
This test port is intended to disturb the PLL just a bit, without causing loss of phase lock, and measure the response. Such work is best done with a dynamic signal analyzer – I’m using a HPAK 3562a, not because it is the latest model, but because that’s what I have around here in my temporary workshop. It had the old CRT replaced by a nice color LCD screen, and it features a very acceptable noise floor, and gain/phases analysis.

The test setup (please excuse the mess, not too much empty bench space around here)
pll loop test - micro-tel sg-811 - adf41020

Now we just need to work through various frequencies and settings, to better understand the characteristics of the system.
To cover all the YIGs and bands of the SG-811 (which might have unknown variations in tuning sensitivity, noise, etc.), frequencies around 2, 6, 10, 12.5 and 17.5 GHz were chosen for the test (exact values can be found in the worksheet, better not to use even values, e.g., 2.0000 GHz, but to exercise the divider circuits – to see if there are any spurs).

At each frequency, magnitude and phase response was collected, examples:
Gain (disregard the unstable response below 10 Hz, just an artifact)
mag_cp0

Phase
phase_cp0

The interesting point is the 0 dB crossing of the gain trace – the unity gain bandwidth. This is determined for each test condition, and then the corresponding phase is obtained from the phase plot. In this example, BW_0dB is about 380 Hz, with about 20 degrees phase. Why is it so important? Simply because we need to keep this phase gap (of the A and B signals) well above 0 degrees, otherwise, the loop will become unstable-oscillate-massive phase noise of the generator will result.

Some call this the phase margin, so do I, although the whole discussion about gain and phase margins is typically centered around open-loop system, whereas we are dealing with a closed loop here. Fair enough.

Now, after some measurements, and number crunching, the results:

Phase vs. BW, at various frequencies
pm vs bw sg-811 pll
-you can see, the phase margin is virtually independent of frequency, and purely a function of bandwidth. So we can limit all further discussion to bandwidth, and don’t need to worry about phase margin separately. It is also clear from this diagram that we should better stay in the 250-300 Hz bandwidth region, for the given filter, to keep the phase margin above 25 degrees, which is a reasonable value.

Now, how to keep the bandwidth stable with all the frequencies and YIGs/SG-811 bands and sensitivities changing? Fortunately, the ADF41020 has a nice build-in function: the charge pump current can be set in 8 steps (0 to 7), from 0.625 to 5 mA (for a 5k1 reference resistor) – and setting the charge pump current (Icp) is not much else than changing the gain of the loop filter. The gain, in turn, will change the 0 dB bandwidth in a fairly linear fashion. Note: typically, the adjustable charge pump current is used to improve locking speed – at wider bandwidth, and mainly, for fixed-frequency applications – but is is also a very useful feature to keep bandwidth stable, for PLL circuits that need to cover a wide range of frequencies, like in the case of the SG-811.

The next result – bandwidth vs. Icp setpoint
sg-811 pll bw vs charge pump current at various frequencies
-looking at this diagram, the bandwidth is not only a function of Icp, but also a function of frequency. For the larger frequencies, the bandwidth is much lower. Some calculations, and it turns out that the product of bandwidth, multiplied with frequency to the power of 0.7 (a bit more than the squareroot) is a good parameter that gives an almost linear vs. Icp (see worksheet, if interested).
adf41020 pll bw phase margin

After all the measurements, things are now pretty clear – if we set the Icp current right, BW can be kept stable, over almost the full range, without any extra parts and switches, and about 300 Hz seems to be a reasonable compromise of PLL speed and stability.

Estimated PLL bandwidth (0 dB), using the Icp current adjustment of the ADF41020
bw vs frq with charge pump current adjustment
At the lowest frequencies (2 GHz range), the BW is found a bit larger than desired, but still, the loop still has 20 degrees margin.

Well, with all the phase margins and uncertainties, is the loop really stable enough? To check this out, what is typically done is to first try a few odd frequencies, at the start, end and in the middle of each band and monitor the VCO control voltage with a scope, for any oscillations or otherwise strange behavior. Then try a few small frequency steps, and see how the loop settles. This all went without any issues.

Still, to be sure, especially close to 2 GHz (increased bandwidth), a test was performed by injecting a 100 mV (nominal) squarewave, 10 Hz, via the test port mentioned above. The loop output spectra showed that this worked, and that the 10 Hz contribution is significant, while still not swamping everything else and driving the loop out of lock right away.

Power spectra with test signal on (upper diagram), and off (lower diagram).
pll power spectra

There are some 60 Hz/harmonic 60 Hz spurs, mainly due to coupling of 60 Hz to the coarse tune line, which is just a plain coax cable that doesn’t provide any good shielding vs. 60 Hz (or 50 Hz, in Europe) interference.

Needless to say, the PLL will not stop working right away when the phase hits 0 deg at the 0 dB point (see above, phase margin vs. bandwidth plot – even at negative phase, measurement was still possible – as long as the amplitude of the test signal is kept small).
There will be signs of instability, and this is what this test reveals. So the frequency was set again to 2.2221 GHz, and the charge pump current Icp increase step by step, from 0 to 5. At 6 and 7, no phase lock could be achieve – fully unstable loop.

Step response (AC component only, square wave, 10 Hz at nominal 100 mV, supplied to test port)
pll step response 2.2221 ghz 100 mV
Icp=0 – this is the most stable condition, phase margin is about 20 degrees. Already at Icp=1, phase margin of about 3 degrees, stability is much compromised/considerably more noise, not only for the step response, but also during the steady portions. At Icp=2 and above, phase margin is negative, still, phase lock is robust (will not re-lock, once lock is lost), and the pulse response suggests to stay away from such regions.

Remotely controlling the Micro-Tel SG-811

The SG-811 comes with various option – mine didn’t come with the IEEE-488 remote control option. At least, it has a BCD type TTL interface. All the essential functions (band, operation mode, attenuators, and in particular, external frequency control-phase lock input enable) can be controlled via no less than 23 signals, plus ground.

All the signals are available at the rear of the instrument, via a 50-pin Centronics connector (similar to the old-fashioned SCSI connectors).

20140825_130155

Several steps were taken to make sure that the ancient but still valuable SG-811 will carefully listen to the commands of a modern area microcontroller:

(1) Fabricate a suitable connector cable. Centronics 50 to D-sub 25. Starting from a pre-assembled D-sub 25 1:1 cable, cut in half, the Centronics connector was soldered on. Quite an effort! Turned out that the 1:1 cable uses pretty thin wire – they are saving on copper, over there, in China!

20140825_124509

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(2) A little shift register, 3×8 bits (3x 74LS164) – a total of 24 wires that can be controlled. 3 of these wires will be used to select the band of the 1295 receiver (via optocouplers, PC817), the reminder, via direct TTL connection, for the SG-811. The shift registers will later be set by a microcontroller, just using 2 outputs to set 24 wires.

20140825_124528

The Microwave PLLs: stabilizing the YIGs

The Micro-Tel SG-811 and 1295 are great units, however, they lack PLL control. Even at their time, in the late 70s, early 80s, government labs required PLL control – and Micro-Tel offered PLL controlled frequency stabilizers for these units. Stabilizers that are now virtually impossible to source (if you have two spare Micro-Tel FS1000, please let me know!).

So I decided to build some very broadband PLL circuits that can handle 2 to 18 GHz, at reasonable frequency resolution. 10 kHz, or 100 kHz resolution seems to be perfectly adequate; mostly, the attenuator calibrator will be used in 2 GHz steps anyway.

Both units have two inputs:

(1) A frequency control input – a voltage controlled input, 0 to 10 V, that sets the frequency roughly, within the given band. Bands are: 2-4, 4-8, 8-12, 12-18 GHz. There is some thermal drift, but preliminary test shows that a 16 bit DAC would be most suitable for this kind of “coarse” frequency control.

(2) A phase lock input. This has a sensitivity of a few MHz per Volt. 0 to 10 V input, for the 1295 – and -3 to 3 V for the SG-811, as it turns out. Accordingly, with the coarse control set to the right value, the phase lock voltage should be somewhere around 3-7 Volts, for the 1295, and close to 0 V for the SG-811.

Now, the tricky part, how to get a phase comparator running, for the 2-18 GHz range? Traditionally, this requires a broadband harmonic generator, locking to a certain harmonic, and so on. All possible, has been done before, but a lot of work to get it working.

There comes the rescue, from Analog Devices: a truely remarkable little thing called ADF41020. It is a full 18 GHz PLL circuit, works with more or less any reference (10 MHz will be used here), and has pretty high input sensitivity, all that is needed are about -10 dBm to drive it over the full band.

After some tricky soldering, in dead-bug style, and some auxilliary circuitry, with 16 bit DAC, reference voltage supply, very clean and stable supplies for the PLL, all the typical loop filters (0.5 KHz bandwidth) – and an ATMega32L – this is the current setup, for the 1295. Believe me, it is working just fine, and even has an auto-track feature, to keep the phase lock voltage mid-range – so it won’t un-lock with drift.

20140825_131843

Upper left hand corner: ADF41020
Lower left hand corner: PLL loop filter
Center: Low noise voltage regulators, reference and DAC
Other parts: ATMega32L board (16 MHz, USB interface), LCD display (just for troubleshooting)

Equipment selection: switching matrix

There are quite a few coaxial switches around – I figured that I need two transfer switches to accomplish the task of “through” calibration, and reflection/insertion loss measurement.
Any unused ports should be automatically terminated with 50 Ohms, when switched out.

Looking around, I found that the HP/Agilent/Keysight (will call it HPAK from now on, and add further letters, with next name change of this wonderful company) HPAK 8763B transfer switch, offers really good data, especially on repeatability. 0.03 dB – for millions of cycles.
Determining this switching reproducibility will be the first task for the attenuation calibrator!

They go for USD 813 each (August 2014), but you can find them much cheaper elsewhere. Preferably, get a unit that doesn’t have 10 million+ cycles yet!

These are of latching type – so we will have to device some drive circuitry to switch them, 24 V positive supply. Won’t be too difficult.

Interconnections will all be rigid coax, and precision SMA to N test cables to connect to source/receiver.

20140825_130253

Note the Sage 0.5-18 GHz coupler, left of the switches. This will be used to get a sample of the SG-811 signal – stay tuned.
For this coupler – this item was found on xbay, quite reasonably prices for its bandwidth. However, the coupled port has a little damage of the SMA connector – rendering it non-usable for its original destiny, but will now be very handy for this project.

To the outside world, the interface is a pair of HPAK SMA (3.5 mm) to precision N panelmount transitions. These are the best and most reliable know to me to date.

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Equipment selection: reflection bridge

Attenuation is defined as insertion loss minus reflection loss.

The insertion loss measurements – that’s quite straightforward, with signal genarator and receiver. We will deal with the particulars later.

For the reflection loss, we still need another device, a directional device. Either a directional coupler, or a return loss bridge.

After careful review, I selected a Narda 5082 “Precision high directivity bridge”. Several reasons:

(1) It is a fairly robust device, and offers N and APC-7 connectors. Luckily, APC-7 adaptors were included. Also included was a combined short/open, APC-7 style. That’s really great.

(2) It is very broadband, 2-18 Ghz full range with one device. This eliminates connections – there are hardly and couplers available that offer 35+ dB directivity, over the full band.

(3) The bridge has a bit more insertion loss compared to a coupler/multi-coupler solution, about 6 dB, but the loss is well defined and flat, will be calibrated out.

(4) It was available, at a few cents for the list price in dollars, and in pristine condition.

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Next step: need to connect the “reflected” port to a switch matrix, via an APC-7 to SMA adapter (which I don’t have in my collection).